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LTC3770 Fast No RSENSETM Step-Down Synchronous Controller with Margining, Tracking and PLL
FEATURES

DESCRIPTIO
Wide VIN Range: 4V to 32V 0.67% 0.6V Reference Voltage Output Voltage Tracking Capability Programmable Margining Sense Resistor Optional True Current Mode Control 2% to 90% Duty Cycle at 200kHz tON(MIN) 100ns Phase Lock Loop Frequency Synchronization Powerful Dual N-Channel MOSFET Driver Adjustable Cycle-by-Cycle Current Limit Adjustable Switching Frequency Programmable Soft-Start Current Foldback Protection (Disabled at Start-Up) Output Overvoltage Protection Micropower Shutdown: IQ < 30A Power Good Output Voltage Monitor Tracks the Reference Input Pin Available in (5mm x 5mm) QFN and 28-Lead SSOP Packages
The LTC(R)3770 is a synchronous step-down switching regulator controller with output voltage up/down tracking capability and voltage margining. Its advanced functions and high accuracy reference are ideal for powering high performance server, ASIC and computer memory systems. The LTC3770 uses a constant on-time, valley current mode control architecture to deliver very low duty factors without requiring a sense resistor. The operating frequency is selected by an external resistor and is compensated for variations in input supply voltage. An internal phase-lock loop allows the IC to be synchronized to an external clock. Fault protection is provided by an overvoltage comparator and input undervoltage lockout. The regulator current limit is user programmable. A wide supply range allows voltages as high as 32V to be stepped down to as low as a 0.6V output. Power supply sequencing is accomplished using an external soft-start timing capacitor.
, LTC and LT are registered trademarks of Linear Technology Corporation. All other trademarks are the property of their respective owners. No RSENSE is a trademark of Linear Technology Corporation. Protected by U.S. Patents including 5481178, 5487554, 6580258, 6304066, 6476589, 6774611.
APPLICATIO S

Distributed Power Systems Server Power Supply
TYPICAL APPLICATIO
0.01F PGOOD MARGIN 10k PLLFLTR PLLIN 0.1F ION VIN TG 0.22F CMDSH-3 68k
High Efficiency Step-Down Converter
VIN 5V TO 28V Si4884 1.8H 10F 35V VOUT x3 2.5V 10A 180F 4V x2
Efficiency and Power Loss vs Load Current
100 VIN = 5V 95 VOUT = 2.5V 90 EFFICIENCY 10
SW TRACK/SS BOOST I
TH
EFFICIENCY (%)
85 80 75 70 65 60 POWER LOSS
+
1000pF
10k
LTC3770 SGND RUN VON VRNG VREFOUT INTVCC DRVCC BG SENSE+
VOUT
Si4874 10F
B340A
95.3k
55 50 0.01 0.1 1 LOAD CURRENT (A) 0.01 10
3770 TA01b
10k VREFIN MPGM 82k
SENSE
-
PGND VFB
3770 TA01
30.1k
U
POWER LOSS (W)
1 0.1
U
U
3770f
1
LTC3770
ABSOLUTE
(Note 1)
AXI U
RATI GS
INTVCC, ZVIN Voltages .................................7V to - 0.3V TG, BG, INTVCC Peak Currents ................................... 4A TG, BG, INTVCC RMS Currents ............................. 50mA Operating Ambient Temperature Range (Note 4) ................................... - 40C to 85C Junction Temperature (Note 2) ............................. 125C Storage Temperature Range ................. - 65C to 125C QFN Reflow Peak Body Temperature .................... 245C Lead Temperature (Soldering, 10 sec).................. 300C
Input Supply Voltage (VIN, VINSNS) ............32V to - 0.3V Boosted Topside Driver Supply Voltage (BOOST) ................................................38V to - 0.3V SENSE+, SW Voltage ....................................32V to - 5V DRVCC, (BOOST - SW) Voltages .................7V to - 0.3V VON, VRNG, PGOOD Voltages .... INTVCC + 0.3V to - 0.3V PLLFLTR, ITH, VFB, VREFIN Voltages ..........2.7V to - 0.3V TRACK/SS, FCB, Z0, Z1, Z2, RUN, PLLIN, MARGIN0, MARGIN1 Voltages ............... INTVCC + 0.3V to - 0.3V
PACKAGE/ORDER I FOR ATIO
TOP VIEW RUN VON PGOOD VRNG VFB ITH SGND MARGIN1 MARGIN0 1 2 3 4 5 6 7 8 9 28 FCB 27 Z0 26 BOOST 25 TG 24 SW 23 PGND 22 BG 21 INTVCC 20 Z2 19 Z1 18 ZVIN 17 VIN 16 PLLIN 15 PLLFLTR
PGOOD
BOOST
ORDER PART NUMBER LTC3770EG
VRNG 1 VFB 2 ITH 3 SGND 4 MARGIN1 5 MARGIN0 6 ION 7 VREFIN 8
RUN
VON
FCB
SW
TG
Z0
G PART MARKING LTC3770EG
ION 10 VREFIN 11 VREFOUT 12 MPGM 13 TRACK/SS 14
MPGM
TRACK/SS
PLLFLTR
PLLIN
G PACKAGE 28-LEAD PLASTIC SSOP
TJMAX = 125C, JA = 130C/ W
UH PACKAGE 32-LEAD (5mm x 5mm) PLASTIC QFN
TJMAX = 125C, JA = 34C/ W EXPOSED PAD IS SGND (PIN 33) MUST BE SOLDERED TO THE PCB
Consult LTC Marketing for parts specified with wider operating temperature ranges.
VREFOUT
VINSNS
ZVIN
VIN
2
U
U
W
WW
U
W
TOP VIEW
ORDER PART NUMBER LTC3770EUH
24 SENSE+ 23 SENSE- 22 PGND 21 BG
32 31 30 29 28 27 26 25
33
20 DRVCC 19 INTVCC 18 Z2 17 Z1
UH PART MARKING 3770
9 10 11 12 13 14 15 16
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LTC3770
ELECTRICAL CHARACTERISTICS
SYMBOL IQ PARAMETER Input DC Supply Current Normal Operation Shutdown Supply Current Feedback Voltage Accuracy (Note 3) Feedback Voltage Line Regulation Feedback Voltage Load Regulation Run Pin On Threshold Soft-Start Charging Current Feedback Pin Input Current Error Amplifier Transconductance Forced Continuous Threshold Forced Continuous Pin Current On-Time Minimum On-Time Minimum Off-Time Maximum Current Sense Threshold VSENSE- - VSENSE+ Minimum Current Sense Threshold VSENSE- - VSENSE+ Output Overvoltage Fault Threshold Offset Undervoltage Lockout Undervoltage Lockout MARGIN0, MARGIN1 Input Thresholds MPGM Pin Voltage TG Driver Pull-Up On Resistance TG Driver Pull-Down On Resistance BG Driver Pull-Up On Resistance BG Driver Pull-Down On Resistance TG Rise Time TG Fall Time BG Rise Time BG Fall Time Main Control Loop
The denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25C. VIN = 15V unless otherwise noted.
CONDITIONS MIN TYP MAX UNITS
1300 30 VREFIN = VREFOUT; ITH = 1.2V (0C to 85C) VREFIN = VREFOUT; ITH = 1.2V VIN = 4V to 30V, ITH = 1.2V (Note 3) ITH = 0.5V to 1.9V (Note 3) VRUN Rising VSS/TRACK = 0V ITH = 1.2V (Note 3) VFCB = 0V ION = -60A, VON = 1.5V ION = -60A, VON = 0V ION = -180A, VON = 0V VRNG = 1V, VFB = VREFIN - 30mV VRNG = 0V, VFB = VREFIN - 30mV VRNG = INTVCC, VFB = VREFIN - 30mV VRNG = 1V, VFB = VREFIN + 30mV VRNG = 0V, VFB = VREFIN + 30mV VRNG = INTVCC, VFB = VREFIN + 30mV 7 VIN Falling VIN Rising

2200 50 0.604 0.606 - 0.3 1.9 -1.7 100 1.6 0.63 -2 290 150 100 400 153 84 308
A A V V %/V % V A nA mS V A ns ns ns ns mV mV mV mV mV mV
VFB VFB(LINEREG) VFB(LOADREG) VRUN ISS/TRACK IFB gm(EA) VFCB IFCB tON tON(MIN) tOFF(MIN) VSENSE(MAX)
0.596 0.594
0.6 0.6 0.002 - 0.05
1 -1.1 -100 1 0.57 210 90
1.5 -1.4 -20 1.3 0.6 -1 250 115 50 250
113 50 228
133 67 268 - 60 - 30 - 120 10 3.2 3.3 1.4 1.18
VSENSE(MIN)
VFB(OV) VIN(UVLO+) VIN(UVLO-) VMGN(TH) VMPGM TG RUP TG RDOWN BG RUP BG RDOWN TG tr TG tf BG tr BG tf
13 3.9 4
% V V V V
TG High TG Low BG High BG Low CLOAD = 3300pF CLOAD = 3300pF CLOAD = 3300pF CLOAD = 3300pF
1.9 1.2 1.9 0.7 20 20 20 20
2.5 2.5 3 1.5
ns ns ns ns
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3
LTC3770
The denotes specifications which apply over the full operating temperature range, otherwise specifications are TA = 25C. VIN = 15V unless otherwise noted.
SYMBOL VINTVCC VLDO(LOADREG) RPLLIN IPLLFLTR PARAMETER Internal VCC Voltage Internal VCC Load Regulation PLLIN Input Resistance Phase Detector Output Current Sink Capability Source Capability PGOOD Upper Threshold PGOOD Lower Threshold PGOOD Hysteresis PGOOD Low Voltage fPLLIN < f0 fPLLIN > f0 VFB Rising VFB Falling VFB Returning IPGOOD = 5mA 7 -7 CONDITIONS 6V < VIN < 30V ICC = 0mA to 20mA
ELECTRICAL CHARACTERISTICS
Internal VCC Regulator
MIN 4.7
TYP 5 - 0.1 50 -15 15 10 - 10 1.5 0.15
MAX 5.3 2
UNITS V % k A A
Phased-Locked Loop
PGOOD Output VFBH VFBL VFB(HYS) VPGL 13 - 13 3 0.4 % % % V
Note 1: Absolute Maximum Ratings are those values beyond which the life of a device may be impaired. Note 2: TJ is calculated from the ambient temperature TA and power dissipation PD as follows: LTC3770EG: TJ = TA + (PD * 130C/W) LTC3770EUH: TJ = TA + (PD * 34C/W)
Note 3: The 3770 is tested in a feedback loop that adjusts VFB to achieve a specified error amplifier output voltage (ITH). For these tests, VREFOUT = VREFIN. Note 4: The LTC3770E is guaranteed to meet performance specifications from 0C to 70C. Specifications over the -40C to 85C operating temperature range are assured by design, characterization and correlation with statistical process controls.
TYPICAL PERFOR A CE CHARACTERISTICS
Current Sense Threshold vs ITH Voltage
300 VRNG = 2V 1.4V 1V
ON-TIME (ns) 1k 10k
CURRENT SENSE THRESHOLD (mV)
200
100
0.7V 0.5V
ON-TIME (ns)
0
-100
-200
0
0.5
1.0 1.5 2.0 ITH VOLTAGE (V)
4
UW
2.5
3770 G01
On-Time vs ION Current
VVON = 0V
On-Time vs VON Voltage
1200 1000 800 600 400 200 IION = 60A
100
3.0
10 1 10 ION CURRENT (A) 100
3770 G02
0
0
1
2 3 VON VOLTAGE (V)
4
5
3770 G03
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LTC3770 TYPICAL PERFOR A CE CHARACTERISTICS
On-Time vs Temperature
MAXIMUM CURRENT SENSE THRESHOLD (mV)
300 250 200 150 100 50 0 -50 -25 IION = 30A VVON = 0V
MAXIMUM CURRENT SENSE THRESHOLD (mV)
ON-TIME (ns)
50 25 75 0 TEMPERATURE (C)
Error Amplifier gm vs Temperature
1.6
SHUTDOWN CURRENT (A)
1.4
INPUT CURRENT (mA)
gm (mS)
1.2
1.0
0.8
0.6 -50
-25
50 25 0 75 TEMPERATURE (C)
INTVCC Load Regulation
0 0 -0.25
UNDERVOLTAGE LOCKOUT THRESHOLD (V)
FCB PIN CURRENT (A)
-0.1
INTVCC (%)
-0.2
-0.3
-0.4
0
10 30 40 20 INTVCC LOAD CURRENT (mA)
UW
100
3770 G04
Maximum Current Sense Threshold vs VRNG Voltage
300 250 200 150 100 50 0
Maximum Current Sense Threshold vs Temperature
150 VRNG = 1V
140
130
120
110
125
0.5
0.75
1.0 1.25 1.5 VRNG VOLTAGE (V)
1.75
2.0
100 -50 -25
50 25 0 75 TEMPERATURE (C)
100
125
3770 G05
3770 G06
Input Current vs Input Voltage
2.5
Shutdown Current vs Input Voltage
60 50 40 30 20 10 0
2.0
1.5
1.0
0.5
0
100 125
0
5
20 15 25 10 INPUT VOLTAGE (V)
30
35
0
5
20 15 25 10 INPUT VOLTAGE (V)
30
35
3770 G07
3770 G08
3770 G09
FCB Pin Current vs Temperature
4.0
Undervoltage Lockout Threshold vs Temperature
3.5
-0.50 -0.75 -1.00 -1.25 -1.50 -50 -25
3.0
2.5
50
3770 G10
50 25 75 0 TEMPERATURE (C)
100
125
2.0 -50 -25
75 0 25 50 TEMPERATURE (C)
100
125
3770 G11
3770 G12
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5
LTC3770 TYPICAL PERFOR A CE CHARACTERISTICS
Track Up
FIGURE 12 CIRCUIT TRACK/SS VFB TRACK/SS AND VFB 500mV/DIV VOUT 2V/DIV 250ms/DIV
3770 G13
Efficiency vs Load Current
100 95 90 DISCONTINUOUS MODE FIGURE 12 CIRCUIT
FREQUENCY (kHz)
ITH VOLTAGE (V)
EFFICIENCY (%)
85 80 75 70 65 60 55 50 0.01 CONTINUOUS MODE
1 0.1 LOAD CURRENT (A)
Efficiency vs Input Voltage
100 95 FCB = 5V FIGURE 12 CIRCUIT 500 450 400 ILOAD = 10A
FREQUENCY (kHz)
EFFICIENCY (%)
90 85 80 75 70 0 5
6
UW
VOUT
10
3770 G16
Track Down
FIGURE 12 CIRCUIT TRACK/SS TRACK/SS AND VFB 500mV/DIV VFB VOUT IL 5A/DIV STEP 0A TO 10A 250ms/DIV
3770 G14
Transient Response
FIGURE 12 CIRCUIT VOUT 100mV/DIV
VOUT 2V/DIV
20s/DIV
3770 G15
ITH Voltage vs Load Current
2.5 FIGURE 12 CIRCUIT 480 460 2.0 440 420 400 380 360 340
Frequency vs Input Voltage
IOUT = 10A
1.5 CONTINUOUS MODE
1.0
IOUT = 0A
0.5
DISCONTINUOUS MODE 0 2 4 6 8 LOAD CURRENT (A) 10 12
3770 G17
0
320 FCB = 0V FIGURE 12 CIRCUIT 300 0 20 5 10 15 INPUT VOLTAGE (V)
25
30
3770 G18
Frequency vs Load Current
FIGURE 12 CIRCUIT
350 300 250 200 150 100 50 0 DISCONTINUOUS MODE CONTINUOUS MODE
ILOAD = 1A
20 10 15 INPUT VOLTAGE (V)
25
30
3770 G19
0
2
4 6 8 LOAD CURRENT (A)
10
12
3770 G20
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LTC3770 TYPICAL PERFOR A CE CHARACTERISTICS
Current Limit Foldback
MAXIMUM CURRENT SENSE THRESHOLD (mV) 160 140 120 100 80 60 40 20 0 0 0.1 0.2 0.3 VFB (V) 0.4 0.5 0.6
3770 G21
VRNG = 1V
ION CURRENT (A)
PI FU CTIO S
(UH Package/G Package)
VRNG (Pin 1/Pin 4): Sense Voltage Range Input. The voltage at this pin is ten times the nominal sense voltage at maximum output current and can be set from 0.5V to 2V by a resistive divider from INTVCC. The nominal sense voltage defaults to 50mV when this pin is tied to ground, 200mV when tied to INTVCC. Do not set this voltage between 0.5V to ground or 2V to INTVCC. VFB (Pin 2/Pin 5): Error Amplifier Feedback Input. This pin connects the error amplifier input to an external resistive divider from VOUT. ITH (Pin 3/Pin 6): Current Control Threshold and Error Amplifier Compensation Point. The current comparator threshold increases with this control voltage. The voltage ranges from 0V to 2.4V with 0.75V corresponding to zero sense voltage (zero current). There is an integrated capacitor of 20pF connected to this pin. SGND (Pin 4/Pin 7): Signal Ground. All small-signal components and compensation components should connect to this ground, which in turn connects to PGND at one point. MARGIN1 (Pin 5/Pin 8): The MSB Logic Input for the Margining Function. Together with the MARGIN0 pin determines whether the IC is in margin high, margin low, or no margin state. This pin has a 50k internal pull-down resistor.
UW
Ion Current vs VIN
140 120 100 80 60 40 20 0 0 5 20 15 25 10 INPUT VOLTAGE (V) 30 35 RON = 82k
3770 G22
U
U
U
MARGIN0 (Pin 6/Pin 9): The LSB Logic Input for the Margining Function. Together with the MARGIN1 pin determines whether the IC is in margin high, margin low, or no margin state. This pin has a 50k internal pull-down resistor. ION (Pin 7/Pin 10): On-Time Current Input. Tie a resistor from this pin to ground to set the one-shot timer current and thereby set the switching frequency. VREFIN (Pin 8/Pin 11): Error Amplifier Reference Input. The voltage at this pin must be greater than 0.5V and less than 1V. VREFOUT (Pin 9/Pin 12): Buffered Internal 0.6V Reference Output. The maximum current sinking limit is 50A at this pin. Do not put a filter capacitor larger than 100pF on this pin. MPGM (Pin 10/Pin 13): Programmable Margining Input. A resistor from this pin to ground sets the margining current. This current, together with the resistor between the VREFOUT and VREFIN pins, determines the margining voltage offset. TRACK/SS (Pin 11/Pin 14): Output Voltage Tracking and Soft Start Input. When the IC is configured to be the master of two outputs, a capacitor to ground at this pin sets the ramp rate for the output voltage. When the IC is configured
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7
LTC3770
PI FU CTIO S (UH Package/G Package)
to be the slave of two outputs, the VFB voltage of the master IC is reproduced by a resistor divider and applied to this pin. An internal 1.4A soft start current is charging this pin during the soft-start phase. PLLFLTR (Pin 12/Pin 15): The Phase-Locked Loop's Lowpass Filter is Tied to This Pin. The voltage at this pin defaults to 1.18V when the IC is not synchronized with an external clock at the PLLIN pin. PLLIN (Pin 13/Pin 16): External Synchronization Input to Phase Detector. This pin is internally terminated to SGND with a 50k resistor. VIN (Pin 14/Pin 17): Main Input Supply. Decouple this pin to PGND with a capacitor (0.1F to 1F). VINSNS (Pin 15) UH Package: VIN Voltage Sense Input. Normally this pin is tied to VIN. However, in certain applications when the IC is powered from a separate supply, VINSNS is tied to the upper MOSFET supply to sense the VIN voltage. The pin is co-bonded with VIN in the SSOP package. ZVIN (Pin 16/Pin 18): Post-Package Zener-Trim Voltage Input. Under normal conditions this pin should always be connected to INTVCC. Z1 (Pin 17/Pin 19): Post-Package Zener-Trim Control. This pin is a multifunctional pin used in production for post-package trimming and tracking. Ground this pin under normal soft-start operation. Connecting this pin to INTVCC will turn off the soft-start current during tracking. Z2 (Pin 18/Pin 20): Post-Package Zener-Trim Control. This pin is used in production for Post-Package trimming. Ground this pin or tie to INTVCC under normal operation. INTVCC (Pin 19/Pin 21): Internal 5V Regulator Output. The control circuits are powered from this voltage. Decouple this pin to PGND with a minimum of 10F low ESR tantalum or ceramic capacitor. DRVCC (Pin 20) UH Package Gate: Driver Voltage Input. Normally connected to the INTVCC regulated output. Do not exceed 7V at this pin. This pin is co-bonded to INTVCC internally in the SSOP package. BG (Pin 21/Pin 22): Bottom Gate Driver Output. This pin drives the gate of the bottom N-channel MOSFET between ground and INTVCC. PGND (Pin 22/Pin 23): Power Ground. Connect this pin closely to the source of the bottom N-channel MOSFET, the (-) terminal of CVCC and the (-) terminal of CIN. SENSE- (Pin 23) UH Package: Current Sense Comparator Input. The (-) input to the current comparator is used to accurately Kelvin sense the bottom side of the sense resistor or MOSFET. This pin is co-bonded with PGND internally in the SSOP package. SENSE+ (Pin 24) UH Package: Current Sense Comparator Input. The (+) input to the current comparator is normally connected to the SW node unless using a sense resistor. This pin is co-bonded with SW internally in the SSOP package. SW (Pin 25/Pin 24): Switch Node. The (-) terminal of the boot-strap capacitor CB connects here. This pin swings from a diode voltage drop below ground up to VIN. TG (Pin 26/Pin 25): Top Gate Drive Output. This pin drives the top N-channel MOSFET with a voltage swing equal to INTVCC, superimposed on the switch node voltage SW. BOOST (Pin 27/Pin 26): Boosted Floating Driver Supply. The (+) terminal of the boot-strap capacitor CB connects here. This pin swings from a diode voltage drop below INTVCC up to VIN + INTVCC. Z0 (Pin 28/Pin 27): Dead Time Control Input. Applying a DC voltage will vary the dead time between TG-Low and BG-High transition. Do not force a voltage higher than 5V on this pin. FCB (Pin 29/Pin 28): Forced Continuous Input. Connect this pin to SGND to force continuous synchronization operation at low load, to INTVCC to enable discontinuous mode operation at low load or to a resistive divider from a secondary output when using a secondary winding. RUN (Pin 30/Pin 1): Run Control Input. A voltage above 1.5V turns on the IC. Forcing this pin below 1.5V shuts down the device.
8
U
U
U
3770f
LTC3770
PI FU CTIO S (UH Package/G Package)
VON (Pin 31/Pin 2): On-Time Voltage Input. Connecting this pin to the output voltage makes the on-time proportional to VOUT. The comparator input defaults to 0.6V when the pin is grounded and defaults to 4.8V when the pin is tied to INTVCC. PGOOD (Pin 32/Pin 3): Power Good Output. Open drain logic output that is pulled to ground when the output voltage is not within 10% of the regulation point, after the internal 25s power bad mask timer expires. Exposed Pad (Pin 33) UH Package: Signal Ground. Must be soldered to the PCB ground for electrical contact and optimum thermal performance.
FU CTIO AL DIAGRA
PLLFLTR 12 VOUT 31 VON 4.8V 0.6V
PLL-SYNC R
PLLIN 13
tON =
VVON (10pF) IION
R4 MPGM 10 1.18V
+
ICMP
-
+
2.0V VRNG 1
-
MARGIN0 6 MARGIN1 5 0.5V
x (0.5~2)
FOLDBACK
VIN 0.6V REF 1 240k Q2 Q4 ITHB Q6 OV
Q1 UV
-
+
EA
80% * VREFIN
1.5V
12K 9 VREFOUT R3 VREFIN 8 3 ITH RC CC1 RUN 30 11 TRACK/SS CSS
+
-
+
-
-++
W
U
U
U
U
U
(UH Package)
RON VIN
7 ION
29 FCB
INTVCC 1A
5V REG
14 VIN
+
CIN
R 15 R VINSNS
ZVIN 0.6V Z0 28 17 18 BOOST 27 CB M1 16 INTVCC 19 Z1 Z2
-
F
+
R S Q FCNT ON SWITCH LOGIC AND ANTISHOOT THROUGH
TG 26 SW 25 SENSE+ 24 SENSE- 23 DRVCC 20 BG 21 CVCC PGND
20k
+
IREV
DB
L1 VOUT
-
RUN
+
COUT M2 RSENSE
(OPTIONAL)*
OV FOLDBACK DISABLED AT START-UP*
0.25V
22 32 PGOOD
+
3.3A
-
+ -
10K VFB 2 R1 R2
- +
- +
SS RUN INTVCC 1.4A
10K
SGND 4 SW SENSE+
90K
BG SENSE-
M2
PGND *CONNECTION W/O SENSE RESISTOR
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LTC3770
OPERATIO
Main Control Loop The LTC3770 is a current mode controller for DC/DC step-down converters. In normal operation, the top MOSFET is turned on for a fixed interval determined by a one-shot timer OST. When the top MOSFET is turned off, the bottom MOSFET is turned on until the current comparator ICMP trips, restarting the one-shot timer and initiating the next cycle. Inductor current is determined by sensing the voltage between the SENSE- (PGND on G Package) and SENSE+ (SW on G Package) pins using a sense resistor or the bottom MOSFET on-resistance . The voltage on the ITH pin sets the comparator threshold corresponding to inductor valley current. The error amplifier EA adjusts this voltage by comparing the feedback signal VFB from a reference voltage set by the VREFIN pin. If the load current increases, it causes a drop in the feedback voltage relative to the reference. The ITH voltage then rises until the average inductor current again matches the load current. At low load currents, the inductor current can drop to zero and become negative. This is detected by current reversal comparator IREV which then shuts off M2, resulting in discontinuous operation. Both switches will remain off with the output capacitor supplying the load current until the ITH voltage rises above the zero current level (0.75V) to initiate another cycle. Discontinuous mode operation is disabled by comparator F when the FCB pin is brought below 0.6V, forcing continuous synchronous operation. The operating frequency is determined implicitly by the top MOSFET on-time and the duty cycle required to maintain regulation. The one-shot timer generates an ontime that is proportional to the ideal duty cycle, thus holding frequency approximately constant with changes in VIN. The nominal frequency can be adjusted with an external resistor RON.
10
U
For applications with stringent constant frequency requirements, the LTC3770 can be synchronized with an external clock. By programming the nominal frequency of the LTC3770 the same as the external clock frequency, the LTC3770 behaves as a constant frequency part against the load and supply variations. Overvoltage and undervoltage comparators OV and UV pull the PGOOD output low if the output feedback voltage exits a 10% window around the regulation point after the internal 25s power bad mask timer expires. Furthermore, in an overvoltage condition, M1 is turned off and M2 is turned on immediately and held on until the overvoltage condition clears. Foldback current limiting is provided if the output is shorted to ground. As VFB drops, the buffered current threshold voltage ITHB is pulled down and clamped to 0.9V. This reduces the inductor valley current level to one tenth of its maximum value as VFB approaches 0V. Foldback current limiting is disabled at start-up. Pulling the RUN pin low forces the controller into its shutdown state, turning off both M1 and M2. Forcing a voltage above 1.5V will turn on the device. INTVCC Power Power for the top and bottom MOSFET drivers and most of the internal controller circuitry is derived from the INTVCC pin. The top MOSFET driver is powered from a floating bootstrap capacitor CB. This capacitor is recharged from INTVCC through an external Schottky diode DB when the top MOSFET is turned off. If the input voltage is low and INTVCC drops below 3.2V, undervoltage lockout circuitry prevents the power switches from turning on.
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LTC3770
APPLICATIO S I FOR ATIO
The basic LTC3770 application circuit is shown in Figure 12. External component selection is primarily determined by the maximum load current and begins with the selection of the sense resistance and power MOSFET switches. The LTC3770 uses either a sense resistor or the on-resistance of the synchronous power MOSFET for determining the inductor current. The desired amount of ripple current and operating frequency largely determines the inductor value. Finally, CIN is selected for its ability to handle the large RMS current into the converter and COUT is chosen with low enough ESR to meet the output voltage ripple and transient specification. Maximum Sense Voltage and VRNG Pin Inductor current is determined by measuring the voltage across a sense resistance that appears between the SENSE- (PGND on G Package) and SENSE+ (SW on G Package) pins. The maximum sense voltage is set by the voltage applied to the VRNG pin and is equal to approximately (0.133)VRNG. The current mode control loop will not allow the inductor current valleys to exceed (0.133)VRNG/RSENSE. In practice, one should allow some margin for variations in the LTC3770 and external component values and a good guide for selecting the sense resistance is:
RSENSE = VRNG 10 * IOUT(MAX)
T NORMALIZED ON-RESISTANCE
An external resistive divider from INTVCC can be used to set the voltage of the VRNG pin between 0.5V and 2V resulting in nominal sense voltages of 50mV to 200mV. Additionally, the VRNG pin can be tied to SGND or INTVCC in which case the nominal sense voltage defaults to 50mV or 200mV, respectively. The maximum allowed sense voltage is about 1.33 times this nominal value. Connecting the SENSE+ and SENSE- Pins The LTC3770 comes in UH and G packages. The UH package IC can be used with or without a sense resistor. When using a sense resistor, place it between the source of the bottom MOSFET, M2, and PGND. Connect the SENSE+ and SENSE- pins to the top and bottom of the sense resistor. Using a sense resistor provides a well defined current limit, but adds cost and reduces efficiency. Alternatively, one can eliminate the sense resistor and use
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the bottom MOSFET as the current sense element by simply connecting the SENSE+ pin to the SW pin and SENSE- pin to PGND. This improves efficiency, but one must carefully choose the MOSFET on-resistance as discussed below. Power MOSFET Selection The LTC3770 requires two external N-channel power MOSFETs, one for the top (main) switch and one for the bottom (synchronous) switch. Important parameters for the power MOSFETs are the breakdown voltage V(BR)DSS, threshold voltage V(GS)TH, on-resistance RDS(ON), reverse transfer capacitance CRSS and maximum current IDS(MAX). The gate drive voltage is set by the 5V INTVCC supply. Consequently, logic-level threshold MOSFETs must be used in LTC3770 applications. If the input voltage is expected to drop below 5V, then sub-logic level threshold MOSFETs should be considered. When the bottom MOSFET is used as the current sense element, particular attention must be paid to its onresistance. MOSFET on-resistance is typically specified with a maximum value RDS(ON)(MAX) at 25C. In this case, additional margin is required to accommodate the rise in MOSFET on-resistance with temperature:
RDS(ON)(MAX) = RSENSE T
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The T term is a normalization factor (unity at 25C) accounting for the significant variation in on-resistance
2.0
1.5
1.0
0.5
0 - 50
50 100 0 JUNCTION TEMPERATURE (C)
150
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Figure 1. RDS(ON) vs Temperature
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with temperature, typically about 0.4%/C as shown in Figure 1. For a maximum junction temperature of 100C, using a value T = 1.3 is reasonable. The power dissipated by the top and bottom MOSFETs strongly depends upon their respective duty cycles and the load current. When the LTC3770 is operating in continuous mode, the duty cycles for the MOSFETs are:
V DTOP = OUT VIN V -V DBOT = IN OUT VIN
The resulting power dissipation in the MOSFETs at maximum output current are: PTOP = DTOP IOUT(MAX)2 T(TOP) RDS(ON)(MAX) + k VIN2 IOUT(MAX) CRSS f PBOT = DBOT IOUT(MAX)2 T(BOT) RDS(ON)(MAX) Both MOSFETs have I2R losses and the top MOSFET includes an additional term for transition losses, which are largest at high input voltages. The constant k = 1.7A-1 can be used to estimate the amount of transition loss. The bottom MOSFET losses are greatest when the bottom duty cycle is near 100%, during a short-circuit or at high input voltage. Operating Frequency The choice of operating frequency is a tradeoff between efficiency and component size. Low frequency operation improves efficiency by reducing MOSFET switching losses but requires larger inductance and/or capacitance in order to maintain low output ripple voltage. The operating frequency of LTC3770 applications is determined implicitly by the one-shot timer that controls the on-time tON of the top MOSFET switch. The on-time is set by the current out of the ION pin and the voltage at the VON pin according to:
V tON = VON (10pF ) IION
SWITCHING FREQUENCY (kHz)
SWITCHING FREQUENCY (kHz)
Tying a resistor RON to SGND from the ION pin yields an on-
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time inversely proportional to 1/3 VIN. The current out of the ION pin is: IION = VIN 3 RON For a step-down converter, this results in approximately constant frequency operation as the input supply varies: f= VOUT [ HZ ] VVON * 3 RON(10pF) To hold frequency constant during output voltage changes, tie the VON pin to VOUT. The VON pin has internal clamps that limit its input to the one-shot timer. If the pin is tied below 0.6V, the input to the one-shot is clamped at 0.6V. Similarly, if the pin is tied above 4.8V, the input is clamped at 4.8V. In high VOUT applications, tie VON to INTVCC. Figures 2a and 2b show how RON relates to switching frequency for several common output voltages.
1000 VOUT = 3.3V VOUT = 2.5V VOUT = 1.5V 100 100 RON (k) 1000
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Figure 2a. Switching Frequency vs RON (VON = 0V)
1000 VOUT = 12V
VOUT = 3.3V
VOUT = 5V
100 10
100 RON (k)
1000
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Figure 2b. Switching Frequency vs RON (VON = INTVCC)
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When there is no RON resistor connected to the ION pin, the on-time tON is theoretically infinite, which in turn could damage the converter. To prevent this, the LTC3770 will detect this fault condition and provide a minimum ION current of 5A to 10A. Changes in the load current magnitude will cause frequency shift. Parasitic resistance in the MOSFET switches and inductor reduce the effective voltage across the inductance, resulting in increased duty cycle as the load current increases. By lengthening the on-time slightly as current increases, constant frequency operation can be maintained. This is accomplished with a resistive divider from the ITH pin to the VON pin and VOUT. The values required will depend on the parasitic resistances in the specific application. A good starting point is to feed about 25% of the voltage change at the ITH pin to the VON pin as shown in Figure 3a. Place capacitance on the VON pin to filter out the ITH variations at the switching frequency. The resistor load on ITH reduces the DC gain of the error amp and degrades load regulation, which can be avoided by using the PNP emitter follower of Figure 3b. Minimum Off-Time and Dropout Operation The minimum off-time tOFF(MIN) is the smallest amount of time that the LTC3770 is capable of turning on the bottom MOSFET, tripping the current comparator and turning the
RVON1 30k VOUT RVON2 100k RC ITH CC CVON 0.01F VON LTC3770
SWITCHING FREQUENCY (MHz)
(3a)
RVON1 3k VOUT 10k INTVCC Q1 2N5087 RVON2 10k CVON 0.01F RC ITH CC
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VON LTC3770
(3b)
Figure 3. Correcting Frequency Shift with Load Current Changes
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MOSFET back off. This time is generally about 250ns. The minimum off-time limit imposes a maximum duty cycle of tON/(tON + tOFF(MIN)). If the maximum duty cycle is reached, due to a dropping input voltage for example, then the output will drop out of regulation. The minimum input voltage to avoid dropout is: VIN(MIN) = VOUT tON + tOFF(MIN) tON A plot of maximum duty cycle vs frequency is shown in Figure 4.
2.0 1.5 DROPOUT REGION 1.0 0.5 0 0 0.25 0.50 0.75 DUTY CYCLE (VOUT/VIN) 1.0
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Figure 4. Maximum Switching Frequency vs Duty Cycle
Inductor Selection Given the desired input and output voltages, the inductor value and operating frequency determine the ripple current: V V IL = OUT 1 - OUT VIN f L Lower ripple current reduces core losses in the inductor, ESR losses in the output capacitors and output voltage ripple. Highest efficiency operation is obtained at low frequency with small ripple current. However, achieving this requires a large inductor. There is a tradeoff between component size, efficiency and operating frequency. A reasonable starting point is to choose a ripple current that is about 40% of IOUT(MAX). The largest ripple current occurs at the highest VIN. To guarantee that ripple current does not exceed a specified maximum, the inductance
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should be chosen according to:
VOUT VOUT L= 1- f IL(MAX) VIN(MAX)
Once the value for L is known, the type of inductor must be selected. High efficiency converters generally cannot afford the core loss found in low cost powdered iron cores, forcing the use of more expensive ferrite, molypermalloy or Kool M(R) cores. A variety of inductors designed for high current, low voltage applications are available from manufacturers such as Sumida, Panasonic, Coiltronics, Coilcraft and Toko. Schottky Diode D1 Selection The Schottky diode D1 shown in Figure 12 conducts during the dead time between the conduction of the power MOSFET switches. It is intended to prevent the body diode of the bottom MOSFET from turning on and storing charge during the dead time, which can cause a modest (about 1%) efficiency loss. The diode can be rated for about one half to one fifth of the full load current since it is on for only a fraction of the duty cycle. In order for the diode to be effective, the inductance between it and the bottom MOSFET must be as small as possible, mandating that these components be placed adjacently. The diode can be omitted if the efficiency loss is tolerable. CIN and COUT Selection The input capacitance CIN is required to filter the square wave current at the drain of the top MOSFET. Use a low ESR capacitor sized to handle the maximum RMS current. IRMS IOUT(MAX) VOUT VIN VIN -1 VOUT
This formula has a maximum at VIN = 2VOUT, where IRMS = IOUT(MAX) / 2. This simple worst-case condition is commonly used for design because even significant deviations do not offer much relief. Note that ripple current ratings from capacitor manufacturers are often based on only 2000 hours of life which makes it advisable to derate the capacitor.
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The selection of COUT is primarily determined by the ESR required to minimize voltage ripple and load step transients. The output ripple VOUT is approximately bounded by:
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1 VOUT IL ESR + 8 fCOUT
Since IL increases with input voltage, the output ripple is highest at maximum input voltage. Typically, once the ESR requirement is satisfied, the capacitance is adequate for filtering and has the necessary RMS current rating. Multiple capacitors placed in parallel may be needed to meet the ESR and RMS current handling requirements. Dry tantalum, special polymer, aluminum electrolytic and ceramic capacitors are all available in surface mount packages. Special polymer capacitors offer very low ESR but have lower capacitance density than other types. Tantalum capacitors have the highest capacitance density but it is important to only use types that have been surge tested for use in switching power supplies. Aluminum electrolytic capacitors have significantly higher ESR, but can be used in cost-sensitive applications providing that consideration is given to ripple current ratings and long term reliability. Ceramic capacitors have excellent low ESR characteristics but can have a high voltage coefficient and audible piezoelectric effects. The high Q of ceramic capacitors with trace inductance can also lead to significant ringing. When used as input capacitors, care must be taken to ensure that ringing from inrush currents and switching does not pose an overvoltage hazard to the power switches and controller. To dampen input voltage transients, add a small 5F to 50F aluminum electrolytic capacitor with an ESR in the range of 0.5 to 2. High performance through-hole capacitors may also be used, but an additional ceramic capacitor in parallel is recommended to reduce the effect of their lead inductance. Top MOSFET Driver Supply (CB, DB) An external bootstrap capacitor CB connected to the BOOST pin supplies the gate drive voltage for the topside MOSFET. This capacitor is charged through diode DB from INTVCC when the switch node is low. When the top MOSFET turns
Kool M is a registered trademark of Magnetics, Inc.
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on, the switch node rises to VIN and the BOOST pin rises to approximately VIN + INTVCC. The boost capacitor needs to store about 100 times the gate charge required by the top MOSFET. In most applications 0.1F to 0.47F, X5R or X7R dielectric capacitor is adequate. Discontinuous Mode Operation and FCB Pin The FCB pin determines whether the bottom MOSFET remains on when current reverses in the inductor. Tying this pin above its 0.6V threshold enables discontinuous operation where the bottom MOSFET turns off when inductor current reverses. The load current at which current reverses and discontinuous operation begins depends on the amplitude of the inductor ripple current and will vary with changes in VIN. Tying the FCB pin below the 0.6V threshold forces continuous synchronous operation, allowing current to reverse at light loads and maintaining high frequency operation. To prevent forcing current back into the main power supply, potentially boosting the input supply to a dangerous voltage level, forced continuous mode of operation is disabled when the TRACK/SS voltage is 20% below the reference voltage during soft-start or tracking up. Forced continuous mode of operation is also disabled when the TRACK/SS voltage is below 0.1V during tracking down operation. During these two periods, the PGOOD signal is forced low. In addition to providing a logic input to force continuous operation, the FCB pin provides a mean to maintain a flyback winding output when the primary is operating in discontinuous mode. The secondary output VOUT2 is normally set as shown in Figure 5 by the turns ratio N of the
+
VIN TG LTC3770 SW R4 FCB R3 SGND BG PGND
3770 F05
VIN CIN 1N4148
+
T1 1:N
*+
VOUT2 COUT2 1F VOUT1 COUT
Figure 5. Secondary Output Loop
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transformer. However, if the controller goes into discontinuous mode and halts switching due to a light primary load current, then VOUT2 will droop. An external resistor divider from VOUT2 to the FCB pin sets a minimum voltage VOUT2(MIN) below which continuous operation is forced until VOUT2 has risen above its minimum. R4 VOUT2(MIN) = 0.6 V 1 + R3 Fault Conditions: Current Limit and Foldback The maximum inductor current is inherently limited in a current mode controller by the maximum sense voltage. In the LTC3770, the maximum sense voltage is controlled by the voltage on the VRNG pin. With valley current control, the maximum sense voltage and the sense resistance determine the maximum allowed inductor valley current. The corresponding output current limit is: ILIMIT = VSNS(MAX) RDS(ON) 1 + IL T 2 The current limit value should be checked to ensure that ILIMIT(MIN) > IOUT(MAX). The minimum value of current limit generally occurs with the largest VIN at the highest ambient temperature, conditions that cause the largest power loss in the converter. Note that it is important to check for self-consistency between the assumed MOSFET junction temperature and the resulting value of ILIMIT which heats the MOSFET switches. Caution should be used when setting the current limit based upon the RDS(ON) of the MOSFETs. The maximum current limit is determined by the minimum MOSFET onresistance. Data sheets typically specify nominal and maximum values for RDS(ON), but not a minimum. A reasonable assumption is that the minimum RDS(ON) lies the same percentage below the typical value as the maximum lies above it. Consult the MOSFET manufacturer for further guidelines. To further limit current in the event of a short circuit to ground, the LTC3770 includes foldback current limiting. If the output falls by more than 60%, then the maximum sense voltage is progressively lowered to about one tenth of its full value.
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INTVCC Regulator
An internal P-channel low dropout regulator produces the 5V supply that powers the drivers and internal circuitry within the LTC3770. The INTVCC pin can supply up to 50mA RMS and must be bypassed to ground with a minimum of 10F low ESR tantalum capacitor or other low ESR capacitor. Good bypassing is necessary to supply the high transient currents required by the MOSFET gate drivers. Applications using large MOSFETs with a high input voltage and high frequency of operation may cause the LTC3770 to exceed its maximum junction temperature rating or RMS current rating. Most of the supply current drives the MOSFET gates. In continuous mode operation, this current is IGATECHG = f(Qg(TOP) + Qg(BOT)). The junction temperature can be estimated from the equations given in Note 2 of the Electrical Characteristics. For example, the LTC3770EG is limited to less than 14mA from a 30V supply: TJ = 70C + (14mA)(30V)(130C/W) = 125C For applications where more current is needed than INTVCC could supply, INTVCC could be driven by an external supply with a voltage higher than 5.3V. However, the INTVCC pin should not exceed its absolute maximum voltage of 7V. External Gate Drive Buffers The LTC3770 drivers are adequate for driving up to about
BOOST Q1 FMMT619 GATE OF M1 Q2 FMMT720 SW INTVCC Q3 FMMT619 GATE OF M2 Q4 FMMT720 PGND
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10 TG
10 BG
Figure 6. Optional External Gate Driver
50nC into MOSFET switches with RMS currents of 50mA. Applications with larger MOSFET switches or operating at frequencies requiring greater RMS currents will benefit from using external gate drive buffers such as the LTC1693. Alternately, the external buffer circuit shown in Figure 6 can be used.
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Soft-Start and Tracking The LTC3770 has the ability to either soft start by itself with a capacitor or track the output of another supply. When the device is configured to soft start by itself, a capacitor should be connected to the TRACK/SS pin. The LTC3770 is put in a low quiescent current shutdown state (IQ < 30A) if the RUN pin voltage is below 1.5V. The TRACK/SS pin is actively pulled to ground in this shutdown state. Once the RUN pin voltage is above 1.5V, the LTC3770 is powered up. A soft-start current of 1.4A then starts to charge the soft-start capacitor CSS. Pin Z1 must be grounded for soft-start operation. Note that soft-start is achieved not by limiting the maximum output current of the controller but by controlling the ramp rate of the output voltage. Current foldback is disabled during this soft-start phase. During the soft-start phase, the LTC3770 is ramping the reference voltage until it is 20% below the voltage set by the VREFIN pin. The force continuous mode is also disabled and PGOOD signal is forced low during this phase. The total soft-start time can be calculated as: tSOFTSTART = 0.8 * VREFIN * CSS/1.4A When the device is configured to track another supply, the feedback voltage of the other supply is duplicated by a resistor divider and applied to the TRACK/SS pin. Pin Z1 should be tied to INTVCC to turn off the soft-start current in this mode. Therefore, the voltage ramp rate on this pin is determined by the ramp rate of the other supply output voltage. Output Voltage Tracking The LTC3770 allows the user to program how its output ramps up and down by means of the TRACK/SS pin. Through this pin, the output can be set up to either coincidentally or ratiometrically track with another supply's output, as shown in Figure 7. In the following discussions, VOUT1 refers to the master LTC3770's output and VOUT2 refers to the slave LTC3770's output. To implement the coincident tracking in Figure 7a, connect an additional resistive divider to VOUT1 and connect its midpoint to the TRACK/SS pin of the slave IC. The ratio of this divider should be selected the same as that of the slave IC's feedback divider shown in Figure 8. In this tracking
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OUTPUT VOLTAGE
VOUT2
OUTPUT VOLTAGE
TIME
(7a) Coincident Tracking
Figure 7. Two Different Modes of Output Voltage Tracking
VOUT1 R3 TO TRACK/SS2 PIN R4 R2 R1 TO VFB1 PIN TO VFB2 PIN R4 R3 TO TRACK/SS2 PIN R2 VOUT2 VOUT1 R1 TO VFB1 PIN TO VFB2 PIN R4
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(8a) Coincident Tracking Setup
Figure 8. Setup for Coincident and Ratiometric Tracking
I
I
+
D1 TRACK/SS2 0.6V VFB2 D3
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D2
EA2
-
Figure 9. Equivalent Input Circuit of Error Amplifier
mode, VOUT1 must be set higher than VOUT2. To implement the ratiometric tracking, the ratio of the divider should be exactly the same as the master IC's feedback divider. Note that the pin Z1 of the slave IC should be tied to INTVCC so that the internal soft-start current is disabled in both tracking modes or it will introduce a small error on the tracking voltage depending on the absolute values of the tracking resistive divider. By selecting different resistors, the LTC3770 can achieve different modes of tracking including the two in Figure 7. So which mode should be programmed? While either mode in Figure 7 satisfies most practical applications,
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VOUT1 VOUT1 VOUT2 TIME
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(7b) Ratiometric Tracking
VOUT2 R3
(8b) Ratiometric Tracking Setup
there do exist some tradeoffs. The ratiometric mode saves a pair of resistors, but the coincident mode offers better output regulation. This can be better understood with the help of Figure 9. At the input stage of the slave IC's error amplifier, two common anode diodes are used to clamp the equivalent reference voltage and an additional diode is used to match the shifted common mode voltage. The top two current sources are of the same amplitude. In the coincident mode, the TRACK/SS voltage is substantially higher than 0.6V at steady state and effectively turns off D1. D2 and D3 will therefore conduct the same current and offer tight matching between VFB2 and the internal precision 0.6V reference. In the ratiometric mode, however, TRACK/SS equals 0.6V at steady state. D1 will divert part of the bias current to make VFB2 slightly lower than 0.6V. Although this error is minimized by the exponential I-V characteristic of the diode, it does impose a finite amount of output voltage deviation. Furthermore, when the master IC's output experiences dynamic excursion (under load transient, for example), the slave IC output will be affected as well. For better output regulation, use the coincident tracking mode instead of ratiometric.
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Margining
Margining is a way to program the reference voltage to the error amplifier to a voltage different from the default 0.6V. Margining is useful for customers who want to stress their systems by varying supply voltages during testing. The reference voltage to the error amplifier is set according to the following equation when the margining function is enabled: VREFIN = 0.6V (1.18V/R4) * R3 Referring to the functional diagram, 0.6V is the buffered system reference at the VREFOUT pin. R3 and R4 are resistors used for programming the amount of margining. VREFIN should be a voltage between 0.5V and 1V. There are two logic control pins, MARGIN1 and MARGIN0, to determine whether the margining function is enabled, Margin up(+) or Margin down(-). Table 1 summarizes the configurations:
Table 1: Margining Function MARGIN1 LOW LOW HIGH HIGH MARGIN0 LOW HIGH LOW HIGH Mode No Margining Margin Up Margin Down No Margining
The buffered reference at VREFOUT has the ability to source a large amount of current. However, it can only sink a maximum of 50A of current. To increase the sinking capability of this reference, connect a resistor to ground at this pin. One may also be tempted to connect a large capacitor to this pin to filter out the noise. However, it is recommended that no larger than 100pF of capacitance should be connected to this pin. Phase-Locked Loop and Frequency Synchronization The LTC3770 has a phase-locked loop comprised of an internal voltage controlled oscillator and phase detector. This allows the top MOSFET turn-on to be locked to the rising edge of an external source. The frequency range of the voltage controlled oscillator is 30% around the center frequency fO. The center frequency is the operating frequency discussed in the previous section. The LTC3770 incorporates a pulse detection circuit that will detect a
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clock on the PLLIN pin. In turn, it will turn on the phaselocked loop function. The pulse width of the clock has to be greater than 400ns and the amplitude of the clock should be greater than 2V. During the start-up phase, phase-locked loop function is disabled. When LTC3770 is not in synchronization mode, PLLFLTR pin voltage is set to around 1.18V. Frequency synchronization is accomplished by changing the internal on-time current according to the voltage on the PLLFLTR pin. The phase detector used is an edge sensitive digital type which provides zero degrees phase shift between the external and internal pulses. This type of phase detector will not lock up on input frequencies close to the harmonics of the VCO center frequency. The PLL hold-in range, fH, is equal to the capture range, fC: fH = fC = 0.3 fO The output of the phase detector is a complementary pair of current sources charging or discharging the external filter network on the PLLFLTR pin. A simplified block diagram is shown in Figure 10. If the external frequency (fPLLIN) is greater than the oscillator frequency fO, current is sourced continuously, pulling up the PLLFLTR pin. When the external frequency is less than fO, current is sunk continuously, pulling down the PLLFLTR pin. If the external and internal frequencies are the same but exhibit a phase difference, the current sources turn on for an amount of time corresponding to the phase difference. Thus the voltage on the PLLFLTR pin is adjusted until the phase and frequency of the external and internal oscillators are identical. At this stable
RLP 2.4V CLP PLLFLTR PLLIN DIGITAL PHASE/ FREQUENCY DETECTOR VCO
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Figure 10. Phase-Locked Loop Block Diagram
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operating point the phase comparator output is open and the filter capacitor CLP holds the voltage. The LTC3770 PLLIN pin must be driven from a low impedance source such as a logic gate located close to the pin. The loop filter components (CLP, RLP) smooth out the current pulses from the phase detector and provide a stable input to the voltage controlled oscillator. The filter components CLP and RLP determine how fast the loop acquires lock. Typically RLP =10k and CLP is 0.01F to 0.1F. Dead Time Control To further optimize the efficiency, the LTC3770 gives users some control over the dead time of the Top gate low and Bottom gate high transition. By applying a DC voltage on the Z0 pin, the TG low BG high dead time can be programmed. Because the dead time is a strong function of the load current and the type of MOSFET used, users need to be careful to optimize the dead time for their particular applications. Figure 11 shows the relation between the TG Low BG High Dead time by varying the Z0 voltages. For an application using LTC3770 with load current of 5A and IR7811W MOSFETs, the dead time could be optimized. To make sure that there is no shoot-through under all conditions, a dead time of 70ns is selected. This corresponds to a DC voltage about 2.6V on Z0 pin. This voltage can easily be generated with a resistor divider off INTVCC.
180 160 140
TDEAD TIME (ns)
120 100 80 60 40 20 0 IOUT = 5A IRT811W FETs 0 1 2 3 Z0 VOLTAGE (V) 4 5
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Figure 11. TG Low BG High Dead Time vs Z0 Voltage
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Efficiency Considerations The percent efficiency of a switching regulator is equal to the output power divided by the input power times 100%. It is often useful to analyze individual losses to determine what is limiting the efficiency and which change would produce the most improvement. Although all dissipative elements in the circuit produce losses, four main sources account for most of the losses in LTC3770 circuits: 1. DC I2R losses. These arise from the resistances of the MOSFETs, inductor and PC board traces and cause the efficiency to drop at high output currents. In continuous mode the average output current flows through L, but is chopped between the top and bottom MOSFETs. If the two MOSFETs have approximately the same RDS(ON), then the resistance of one MOSFET can simply be summed with the resistances of L and the board traces to obtain the DC I2R loss. For example, if RDS(ON) = 0.01 and RL = 0.005, the loss will range from 15mW to 1.5W as the output current varies from 1A to 10A. 2. Transition loss. This loss arises from the brief amount of time the top MOSFET spends in the saturated region during switch node transitions. It depends upon the input voltage, load current, driver strength and MOSFET capacitance, among other factors. The loss is significant at input voltages above 20V and can be estimated from: Transition Loss (1.7A-1) VIN2 IOUT CRSS f 3. INTVCC current. This is the sum of the MOSFET driver and control currents. 4. CIN loss. The input capacitor has the difficult job of filtering the large RMS input current to the regulator. It must have a very low ESR to minimize the AC I2R loss and sufficient capacitance to prevent the RMS current from causing additional upstream losses in fuses or batteries. Other losses, including COUT ESR loss, Schottky diode D1 conduction loss during dead time and inductor core loss generally account for less than 2% additional loss. When making adjustments to improve efficiency, the input current is the best indicator of changes in efficiency.
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If you make a change and the input current decreases, then the efficiency has increased. If there is no change in input current, then there is no change in efficiency. Checking Transient Response The regulator loop response can be checked by looking at the load transient response. Switching regulators take several cycles to respond to a step in load current. When a load step occurs, VOUT immediately shifts by an amount equal to ILOAD (ESR), where ESR is the effective series resistance of COUT. ILOAD also begins to charge or discharge COUT generating a feedback error signal used by the regulator to return VOUT to its steady-state value. During this recovery time, VOUT can be monitored for overshoot or ringing that would indicate a stability problem. The ITH pin external components shown in Figure 12 will provide adequate compensation for most applications. For a detailed explanation of switching control loop theory see Application Note 76. Design Example As a design example, take a supply with the following specifications: VIN = 5V to 28V (15V nominal), VOUT = 2.5V 5%, IOUT(MAX) = 10A, f = 450kHz. First, calculate the timing resistor with VON = VOUT:
RON = 2.5V = 74k 3(2.5V )(450kHz)(10pF )
and choose the inductor for about 40% ripple current at the maximum VIN:
L=
2.5V 2.5V 1- = 1.3H (450kHz)(0.4)(10A) 28V
Selecting a standard value of 1.8H results in a maximum ripple current of:
IL =
2.5V 2.5V 1- = 2.8 A (450kHz)(1.8H) 28V
Next, choose the synchronous MOSFET switch. Choosing a Si4874 (RDS(ON) = 0.0083 (NOM) 0.010 (MAX), JA = 40C/W) yields a nominal sense voltage of: VSNS(NOM) = (10A)(1.3)(0.0083) = 108mV
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Tying VRNG to 1.1V will set the current sense voltage range for a nominal value of 110mV with current limit occurring at 146mV. To check if the current limit is acceptable, assume a junction temperature of about 80C above a 70C ambient with 150C = 1.5:
ILIMIT 146mV 1 + (2.8 A ) = 11A (1.5)(0.010) 2
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and double check the assumed TJ in the MOSFET: PBOT = 28 V - 2 .5V (11A )2 (1.5 )(0.010 ) = 1.65 W 28 V
TJ = 70C + (1.65W)(40C/W) = 136C Because the top MOSFET is on for such a short time, an Si4884 RDS(ON)(MAX) = 0.0165, CRSS = 100pF, JA = 40C/W will be sufficient. Checking its power dissipation at current limit with 100C = 1.4:
PTOP =
2.5V (11A)2 (1.4)(0.0165) + 28 V
(1.7)(28V)2 (11A)(100pF )(250kHz)
= 0.25W + 0.37W = 0.62W
TJ = 70C + (0.62W)(40C/W) = 95C The junction temperature will be significantly less at nominal current, but this analysis shows that careful attention to heat sinking on the board will be necessary in this circuit. CIN is chosen for an RMS current rating of about 3A at 85C. The output capacitors are chosen for a low ESR of 0.013 to minimize output voltage changes due to inductor ripple current and load steps. The ripple voltage will be only: VOUT(RIPPLE) = IL(MAX) (ESR) = (2.8A) (0.013) = 36mV However, a 0A to 10A load step will cause an output change of up to: VOUT(STEP) = ILOAD (ESR) = (10A) (0.013) = 130mV An optional 22F ceramic output capacitor is included to minimize the effect of ESL in the output ripple. The complete circuit is shown in Figure 12.
3770f
LTC3770
APPLICATIO S I FOR ATIO
INTVCC 5V RPG 100k R7 47k R8 51k RUN 1 2 3 4 5 R6 11k 6 7 8 9 10 RC 20k CC2 100pF CC1 500pF 11 R3 10k RON 75k 12 13 14 RUN VON PGOOD VRNG VFB ITH SGND MARGIN1 MARGIN0 ION VREFIN VREFOUT MPGM TRACK/SS LTC3770EG
R5 39k
BOOST TG SW PGND BG INTVCC Z1 Z2 ZVIN VIN PLLIN PLLFLTR
CSS 0.1F R4 82k
R1 30.1k
R2 95.3k
Figure 12. Design Example: 2.5V/10A at 450kHz
To set a 25% margining, select the resistors R3, R4 such that VREFIN = 0.6 25% * 0.6 or 1.18 * R3 = 25% * 0.6 R4 R4 8R3 Choose R3 to be 10k, R4 to be 82k for this application. PC Board Layout Checklist When laying out a PC board follow one of two suggested approaches. The simple PC board layout requires a dedicated ground plane layer. Also, for higher currents, it is recommended to use a multilayer board to help with heat sinking power components. * The ground plane layer should not have any traces and
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FCB Z0 28 27 26 25 24 23 22 21 20 19 18 17 16 15 CVIN 0.1F M2 Si4874 CVCC 10F COUT3 23F x5R x2 DB CMDSH-3 D1 B340A L1 1.8H VOUT 2.5V 10A M1 Si4884 VIN 5V TO 28V CIN 10F 50V x3
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CB 0.22F
+
COUT1-2 180F 4V x2
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L1: SUMIDA CEP125-1R8MC-H COUT: CORNELL DUBILIER ESRE181E04B CIN: UNITED CHEMICON THCR60E1H106ZT
3770 F12
it should be as close as possible to the layer with power MOSFETs. * Place CIN, COUT, MOSFETs, D1 and inductor all in one compact area. It may help to have some components on the bottom side of the board. * Use an immediate via to connect the components to ground plane including SGND and PGND of LTC3770. Use several bigger vias for power components. * Use compact plane for switch node (SW) to improve cooling of the MOSFETs and to keep EMI down. * Use planes for VIN and VOUT to maintain good voltage filtering and to keep power losses low. * Flood all unused areas on all layers with copper. Flooding with copper will reduce the temperature rise of power component. You can connect the copper areas to any DC net (VIN, VOUT, GND or to any other DC rail in your system).
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LTC3770
APPLICATIO S I FOR ATIO
When laying out a printed circuit board, without a ground plane, use the following checklist to ensure proper operation of the controller. * Segregate the signal and power grounds. All small signal components should return to the SGND pin at one point which is then tied to the PGND pin close to the source of M2. * Place M2 as close to the controller as possible, keeping the PGND, BG and SW traces short. * Connect the input capacitor(s) CIN close to the power
PACKAGE DESCRIPTIO
G Package 28-Lead Plastic SSOP (5.3mm)
(Reference LTC DWG # 05-08-1640)
9.90 - 10.50* (.390 - .413) 28 27 26 25 24 23 22 21 20 19 18 17 16 15
7.8 - 8.2
0.42 0.03 RECOMMENDED SOLDER PAD LAYOUT 5.00 - 5.60** (.197 - .221)
0.09 - 0.25 (.0035 - .010)
0.55 - 0.95 (.022 - .037)
NOTE: 1. CONTROLLING DIMENSION: MILLIMETERS MILLIMETERS 2. DIMENSIONS ARE IN (INCHES) 3. DRAWING NOT TO SCALE *DIMENSIONS DO NOT INCLUDE MOLD FLASH. MOLD FLASH SHALL NOT EXCEED .152mm (.006") PER SIDE **DIMENSIONS DO NOT INCLUDE INTERLEAD FLASH. INTERLEAD FLASH SHALL NOT EXCEED .254mm (.010") PER SIDE
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MOSFETs. This capacitor carries the MOSFET AC current. * Keep the high dV/dt SW, BOOST and TG nodes away from sensitive small-signal nodes. * Connect the INTVCC decoupling capacitor CVCC closely to the INTVCC and PGND pins. * Connect the top driver boost capacitor CB closely to the BOOST and SW pins. * Connect the VIN pin decoupling capacitor CF closely to the VIN and PGND pins.
1.25 0.12 5.3 - 5.7 7.40 - 8.20 (.291 - .323) 0.65 BSC 1 2 3 4 5 6 7 8 9 10 11 12 13 14 2.0 (.079) MAX
0 - 8
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0.65 (.0256) BSC
0.22 - 0.38 (.009 - .015) TYP
0.05 (.002) MIN
G28 SSOP 0204
3770f
LTC3770
PACKAGE DESCRIPTIO
5.50 0.05
0.70 0.05
3.45 0.05 (4 SIDES)
4.10 0.05 PACKAGE OUTLINE 0.25 0.05 0.50 BSC RECOMMENDED SOLDER PAD LAYOUT 0.200 REF NOTE: 1. DRAWING PROPOSED TO BE A JEDEC PACKAGE OUTLINE M0-220 VARIATION WHHD-(X) (TO BE APPROVED) 2. DRAWING NOT TO SCALE 3. ALL DIMENSIONS ARE IN MILLIMETERS 4. DIMENSIONS OF EXPOSED PAD ON BOTTOM OF PACKAGE DO NOT INCLUDE MOLD FLASH. MOLD FLASH, IF PRESENT, SHALL NOT EXCEED 0.20mm ON ANY SIDE 5. EXPOSED PAD SHALL BE SOLDER PLATED 6. SHADED AREA IS ONLY A REFERENCE FOR PIN 1 LOCATION ON THE TOP AND BOTTOM OF PACKAGE 0.25 0.05 0.50 BSC
TYPICAL APPLICATIO
RPG 100k PGOOD RUN R7 47k R8 51k 1 2 3 4 R2 60.4k RC 10k MARGIN1 MARGIN0 CC2 100pF CC1 1000pF 5 6 7 RON 75k R5 10k 8 VRNG VFB ITH SGND
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PGOOD VON RUN FCB
RRUN 51k
MARGIN1 MARGIN0 ION VREFIN
CF 220pF
R1 30.1k
VREFOUT MPGM TRACK/SS PLLFLTR PLLIN VIN VINSNS ZVIN 9 10 11 12 13 14 15 16
TRACK/SS R6 200k VCC 5V
Information furnished by Linear Technology Corporation is believed to be accurate and reliable. However, no responsibility is assumed for its use. Linear Technology Corporation makes no representation that the interconnection of its circuits as described herein will not infringe on existing patent rights.
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UH Package 32-Lead Plastic QFN (5mm x 5mm)
(Reference LTC DWG # 05-08-1693)
BOTTOM VIEW--EXPOSED PAD 5.00 0.10 (4 SIDES) 0.75 0.05 0.00 - 0.05 R = 0.115 TYP 31 32 0.40 0.10 1 2 3.45 0.10 (4-SIDES) 0.23 TYP (4 SIDES) PIN 1 TOP MARK (NOTE 6)
(UH) QFN 0603
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1.8V/5A at 450kHz with Tracking
L1: BI TECH 1.8H HM65-H1R8-TB M1, M2: PHILIPS PH3230 CIN: TDK C4532X5R1H685M COUT: PANASONIC EEFUE0G181R FCB VIN 4V TO 28V 28 27 26 25 SW M1 PH3230 31 30 29 Z0 BOOST TG SENSE+ SENSE- PGND BG LTC3770EUH DRVCC INTVCC Z2 Z1 24 23 22 21 20 19 18 17 M2 PH3230 INTVCC DB CMDSH-3 D1 B340A 1000pF CB 0.22F L1 1.8H VOUT 1.8V 5A
+
COUT 180F 4V x2 CIN 6.8F 50V x3
3770 TA02a
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CVCC 10F
COUT3 22F X5R x2
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LTC3770
TYPICAL APPLICATIO
RPG 100k
PGOOD RUN R7 47k 32 R4 39k RRUN 51k R8 51k 1 2 R3 11k 3 4 R2 95.3k RC 10k MARGIN1 MARGIN0 CC2 100pF R1 30.1k CC1 1000pF RON 75k R5 10k 5 6 7 8 VRNG VFB ITH SGND 31 30
PGOOD VON RUN FCB
MARGIN1 MARGIN0 ION VREFIN
CF 220pF
VREFOUT MPGM TRACK/SS PLLFLTR PLLIN VIN VINSNS ZVIN 9 10 11 RPL 10k R6 82k CSS 0.1F CPL 0.01F 12 13 PLLIN 0.1F CP 1000pF CVIN 0.01F RVIN 10 14 15 16
RELATED PARTS
PART NUMBER LTC1622 LTC1625/LTC1775 LTC1628/LTC3728 LTC1735 LTC1736 LTC1772 LTC1773 LTC1778 LTC1876 LTC3708 LTC3713 LTC3731 LTC3778 DESCRIPTION 550kHz Step-Down Controller No RSENSE Current Mode Synchronous Step-Down Controller Dual, 2-Phase Synchronous Step-Down Controller High Efficiency, Synchronous Step-Down Controller High Efficiency, Synchronous Step-Down Controller with 5-Bit VID SOT-23 Step-Down Controller Synchronous Step-Down Controller Wide Range, No RSENSE Synchronous Step-Down Controller 2-Phase, Dual Synchronous Step-Down Controller with Step-Up Regulator Dual, 2-Phase, No RSENSE Synchronous Step-Down Controller with Output Tracking Low VIN High Current Synchronous Step-Down Controller 3-Phase Synchronous Step-Down Controller Low VOUT, No RSENSE Synchronous Step-Down Controller COMMENTS 8-Pin MSOP; Synchronizable; Soft-Start; Current Mode 97% Efficiency; No Sense Resistor; 16-Pin SSOP Power Good Output; Minimum Input/Output Capacitors; 3.5V VIN 36V Burst Mode(R) Operation; 16-Pin Narrow SSOP; 3.5V VIN 36V Mobile VID; 0.925V VOUT 2V; 3.5V VIN 36V Current Mode; 550kHz; Very Small Solution Size Up to 95% Efficiency, 550kHz, 2.65V VIN 8.5V, 0.8V VOUT VIN, Synchronizable to 750kHz GN16-Pin, 0.8VFB Reference 3.5V VIN 36V, Power Good Output, 300kHz Operation Fast Transient Response Reduces COUT; 4V VIN 36V, 0.6V VOUT 6V; 2-Phase Operation Reduces CIN 1.5V VIN 36V, 0.8V VOUT (0.9)VIN, IOUT Up to 20A 600kHz; Up to 60A Output 0.6V VOUT (0.9)VIN, 4V VIN 36V, IOUT Up to 20A
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Burst Mode is a registered trademark of Linear Technology Corporation.
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Linear Technology Corporation
1630 McCarthy Blvd., Milpitas, CA 95035-7417
(408) 432-1900 FAX: (408) 434-0507
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Typical Application 2.5V/10A Synchronized at 450kHz
L1: BI TECH 1.8H HM65-H1R8-TB M1, M2: PHILIPS PH3230 CIN: TDK C4532X5R1H685M COUT: PANASONIC EEFUE0G181R FCB VIN 5V TO 28V 28 27 26 25 SW M1 PH3230 29 Z0 BOOST TG SENSE+ SENSE- PGND BG LTC3770EUH DRVCC INTVCC Z2 Z1 24 23 22 21 20 19 18 17 M2 PH3230 INTVCC CVCC 10F COUT3 22F X5R x2 DB CMDSH-3 D1 B340A 1000pF CB 0.22F L1 1.8H VOUT 2.5V 10A
+
COUT 180F 4V x2 CIN 10F 50V x3
3770 TA02b
LT/TP 1104 1K * PRINTED IN USA
www.linear.com
(c) LINEAR TECHNOLOGY CORPORATION 2004

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